Introduction The SR620 Repairing the SR620 Replacing the Backup 3V Lithium Battery Switching to an External Reference Clock Running Auto-Calibration Oscilloscope Display Mode References Footnotes Introduction A little over a year ago, I found a Stanford Research Systems SR620 universal time interval counter at the Silicon Valley Electronics Flea Market. It had a big sticker “Passes Self-Test” and “Tested 3/9/24” (the day before the flea market) on it so I took the gamble and spent an ungodly $4001 on it. Luckily, it did work fine, initially at least, but I soon discovered that it sometimes got into some weird behavior after pressing the power-on switch. The SR620 The SR620 was designed sometime in the mid-1980s. Mine has a rev C PCB with a date of July 1988, 37 year old! The manual lists 1989, 2006, 2019 and 2025 revisions. I don’t know if there were any major changes along the way, but I doubt it. It’s still for sale on the SRS website, starting at $5150. The specifications are still pretty decent, especially for a hobbyist: 25 ps single shot time resolution 1.3 GHz frequency range 11-digit resolution over a 1 s measurement interval The SR620 is not perfect, one notable issue is its thermal design. It simply doesn’t have enough ventilation holes, the heat-generating power regulators are located close to the high precision time-to-analog converters, and the temperature sensor for the fan is inexplicably placed right next to the fan, which is not close at all to the power regulators. The Signal Path has an SR620 repair video that talks about this. Repairing the SR620 You can see the power-on behavior in the video below: Of note is that lightly touching the power button changes the behavior and sometimes makes it get all the way through the power-on sequence. This made me hopeful that the switch itself was bad, something that should be easy to fix. Unlike my still broken SRS DG535, another flea market buy with the most cursed assembly, the SR620 is a dream to work on: 4 side screws is all it takes to remove the top of the case and have access to all the components from the top. Another 4 screws to remove the bottom panel and you have access to the solder side of the PCB. You can desolder components without lifting the PCB out of the enclosure. Like my HP 5370A, the power switch of the SR620 selects between power on and standby mode. The SR620 enables the 15V rail at all times to keep a local TCXO or OCXO warmed up. The power switch is located at the right of the front panel. It has 2 black and 2 red wires. When the unit is powered on, the 2 black wires and the 2 red wires are connected to each other. To make sure that the switch itself was the problem, I soldered the wires together to create a permanent connection: After this, the SR620 worked totall fine! Let’s replace the switch. Unscrew 4 more screws and pull the knobs of the 3 front potentiometers and power switch to get rid of the front panel: A handful of additional screws to remove the front PCB from the chassis, and you have access to the switch: The switch is an ITT Schadow NE15 T70. Unsurprisingly, these are not produced anymore, but you can still find them on eBay. I paid $7.5 + shipping, the price increased to $9.5 immediately after that. According to this EEVblog forum post, this switch on Digikey is a suitable replacement, but I didn’t try it. The old switch (bottom) has 6 contact points vs only 4 of the new one (top), but that wasn’t an issue since only 4 were used. Both switches also have a metal screw plate, but they were oriented differently. However, you can easily reconfigure the screw plate by straightening 4 metal prongs. If you buy the new switch from Digikey and it doesn’t come with the metal screw plate, you should be able to transplant the plate from the broken switch to the new one just the same. To get the switch through the narrow hole of the case, you need to cut off the pins on the one side of the switch and you need to bend the contact points a bit. After soldering the wires back in place, the SR620 powered on reliably. Switch replacement completed! Replacing the Backup 3V Lithium Battery The SR620 has a simple microcontroller system consists of a Z8800 CPU, 64 KB of EPROM and a 32 KB SRAM. In addition to program data, the SRAM also contains calibration and settings. replaced one such battery in my HP 3478A multimeter. These batteries last almost forever, but mine had a 1987 date code and 38 years is really pushing things, so I replaced it with this new one from Digikey. The 1987 version of this battery had 1 pin on each side, on the new ones, the + side has 2 pins, so you need to cut one of those pins and install the battery slightly crooked back onto the PCB. When you first power up the SR620 after replacing the battery, you might see “Test Error 3” on the display. According to the manual: Test error 3 is usually “self-healing”. The instrument settings will be returned to their default values and factory calibration data will be recalled from ROM. Test Error 3 will recur if the Lithium battery or RAM is defective. After power cycling the device again, the test error was gone and everything worked, but with a precision that was slightly lower than before: before the battery replacement, when feeding the 10 MHz output reference clock into channel A and measuring frequency with a 1s gate time, I’d get a read-out of 10,000,000.000N MHz. In other words: around a milli-Hz accuracy. After the replacment, the accuracy was about an order of magnitude worse. That’s just not acceptable! The reason for this loss in accuracy is because the auto-calibration parameters were lost. Luckily, this is easy to fix. Switching to an External Reference Clock My SR620 has the cheaper TCXO option which gives frequency measurement results that are about one order of magnitude less accurate than using an external OCXO based reference clock. So I always switch to an external reference clock. The SR620 doesn’t do that automatically, you need to manually change it in the settings, as follows: SET -> “ctrl cal out scn” SEL -> “ctrl cal out scn” SET -> “auto cal” SET -> “cloc source int” Scale Down arrow -> “cloc source rear” SET -> “cloc Fr 10000000” SET If you have a 5 MHz reference clock, use the down or up arrow to switch between 1000000 and 5000000. Running Auto-Calibration You can rerun auto-calibration manually from the front panel without opening up the device with this sequence: SET -> “ctrl cal out scn” SEL -> “ctrl cal out scn” SET -> “auto cal” START The auto-calibration will take around 2 minutes. Only run it once the device has been running for a while to make sure all components have warmed up and are at stable temperature. The manual recommends a 30 minute warmup time. After doing auto-calibration, feeding back the reference clock into channel A and measuring frequency with a 1 s gate time gave me a result that oscillated around 10 MHz, with the mHz digits always 000 or 999.2 It’s possible to fine-tune the SR620 beyond the auto-calibration settings. One reason why one might want to do this is to correct for drift of the internal oscillator To enable this kind of tuning, you need to move a jumper inside the case. The time-nuts email list has a couple of discussions about this, here is one such post. Page 69 of the SR620 manual has detailed calibration instructions. Oscilloscope Display Mode When the 16 7-segment LEDs on the front panel are just not enough, the SR620 has this interesting way of (ab)using an oscilloscope as general display: it uses XY mode to paint the data. I had tried this mode in the past with my Sigilent digital oscilloscope, but the result was unreadable: for this kind of rendering, having a CRT beam that lights up all the phosphor from one point to the next is a feature, not a bug. This time, I tried it with an old school analog oscilloscope3: (Click to enlarge) The result is much better on the analog scope, but still very hard to read. When you really need all the data you can get from the SR620, just use the GPIB or RS232 interface. References The Signal Path - TNP #41 - Stanford Research SR620 Universal Time Interval Counter Teardown, Repair & Experiments Some calibration info about the SR620 Fast High Precision Set-up of SR 620 Counter The rest of this page has a bunch of other interesting SR620 related comments. Time-Nuts topics The SR620 is mentioned in tons of threads on the time-nuts emaiml list. Here are just a few interesting posts: This post talks about some thermal design mistakes in the SR620. E.g. the linear regulators and heat sink are placed right next to the the TCXO. It also talks about the location of the thermistor inside the fan path, resulting in unstable behavior. This is something Shrirar of The Signal Path fixed by moving the thermistor. This comment mentions that while the TXCO stays powered on in standby, the DAC that sets the control voltage does not, which results in an additional settling time after powering up. General recommendation is to use an external 10 MHz clock reference. This comment talks about warm-up time needed depending on the desired accuracy. It also has some graphs. Footnotes This time, the gamble paid off, and the going rate of a good second hand SR620 is quite a bit higher. But I don’t think I’ll ever do this again! ↩ In other words, when fed with the same 10 MHz as the reference clock, the display always shows a number that is either 10,000,000,000x or 9,999,999,xx. ↩ I find it amazing that this scope was calibrated as recently as April 2023. ↩

Introduction Finding the Display Tuning Potentiometers The Result Hardcopy Preview Mode Introduction Less than a week after finishing my TDS 684B analog memory blog post, a TDS 684C landed on my lab bench with a very dim CRT. If you follow the lives the 3-digit TDS oscilloscope series, you probably know that this is normally a bit of death sentence of the CRT: after years of use, the cathode ray loses its strength and there’s nothing you can do about it other than replace the CRT with an LCD screen. I was totally ready to go that route, and if I ever need to do it, here are 3 possible LCD upgrade options that I list for later reference: The most common one is to buy a $350 Newscope-T1 LCD display kit by SimmConn Labs. A cheaper hobbyist alternative is to hack something together with a VGA to LVDS interface board and some generic LCD panel, as described in this build report. He uses a VGA LCD Controller Board KYV-N2 V2 with a 7” A070SN02 LCD panel. As I write this, the cost is $75, but I assume this used to be a lot cheaper before tariffs were in place. If you really want to go hard-core, you could make your own interface board with an FPGA that snoops the RAMDAC digital signals and converts them to LVDS, just like the Newscope-T1. There is a whole thread about this the EEVblog forum. But this blog post is not about installing an LCD panel! Before going that route, you should try to increase the brightness of the CRT by turning a potentiometer on the display board. It sounds like an obvious thing to try, but didn’t a lot of reference to online. And in my case, it just worked. Finding the Display Tuning Potentiometers In the Display Assembly Adjustment section of chapter 5 of the TDS 500D, TDS 600C, TDS 700D and TDS 714L Service Manual, page 5-23, you’ll find the instructions on how to change rotation, brightness and contrast. It says to remove the cabinet and then turn some potentiometer, but I just couldn’t find them! They’re supposed to be next to the fan. Somewhere around there: Well, I couldn’t see any. It’s only the next day, when I was ready to take the whole thing apart that I noticed these dust covered holes: A few minutes and a vaccum cleaning operation later reveals 5 glorious potentiometers: From left to right: horizontal position rotation vertical position brightness contrast Rotate the last 2 at will and if you’re lucky, your dim CRT will look brand new again. It did for me! The Result The weird colors in the picture above is a photography artifact that’s caused by Tektronix NuColor display technology: it uses a monochrome CRT with an R/G/B shutter in front of it. You can read more about it in this Hackaday article. In real life, the image looks perfectly fine! Hardcopy Preview Mode If dialing up the brightness doesn’t work and you don’t want to spend money on an LCD upgrade, there is the option of switching the display to Hardcopy mode, like this: [Display] -> [Settings <Color>] -> [Palette] -> [Hardcopy preview] Instead of a black, you will now get a white background. It made the scope usable before I made the brightness adjustment.

Introduction The TDS600 Series The Acquisition Board Measuring Along the Signal Path A Closer Look at the Noise Issue Conclusion Introduction I have a Tektronix TDS 684B oscilloscope that I bought cheaply at an auction. It has 4 channels, 1 GHz of BW and a sample rate of 5 Gsps. Those are respectable numbers even by today’s standards. It’s also the main reason why I have it: compared to modern oscilloscopes, the other features aren’t nearly as impressive. It can only record 15k samples per channel at a time, for example. But at least the sample rate doesn’t go down when you increase the number of recording channels: it’s 5 Gsps even all 4 channels are enabled. I’ve always wondered how Tektronix managed to reach such high specifications back in the nineties, so in this blog post I take a quick look at the internals, figure out how it works, and do some measurements along the signal path. The TDS600 Series The first oscilloscopes of the TDS600 series were introduced around 1993. The last one, the TDS694C was released in 2002. The TDS684 version was from sometime 1995. The ICs on my TDS684C have date codes from as early as the first half of 1997. The main characteristic of these scopes was their extreme sample rate for that era, going from 2 Gsps for the TDS620, TDS640 and TDS644, 5 Gsps for the TDS654, TDS680 and TDS684, and 10 Gsps for the TDS694C which was developed under the Screamer code name. The oscilloscopes have 2 main boards: the acquisition board contains all the parts from the analog input down to the sample memory as well as some triggering logic. (Click to enlarge) a very busy CPU board does the rest. (Click to enlarge) 2 flat cables and a PCB connect the 2 boards. The interconnect PCB traces go to the memory on the acquisition board. It’s safe to assume that this interface is used for high-speed waveform data transfer while the flat cables are for lower speed configuration and status traffic. If you ever remove the interconnection PCB, make sure to put it back with the same orientation. It will fit just fine when rotated 180 degrees but the scope won’t work anymore! The Acquisition Board The TDS 684B has 4 identical channels that can easily be identified. (Click to enlarge) There are 6 major components in the path from input to memory: Analog front-end Hidden under a shielding cover, but you’d expect to find a bunch of relays there to switch between different configurations: AC/DC, 1Meg/50 Ohm termination, … I didn’t open it because it requires disassembling pretty much the whole scope. Signal Conditioner IC(?) This is the device with the glued-on heatsink. I left it in place because there’s no metal attachment latch. Reattaching it would be a pain. Since the acquisition board has a bunch of custom ICs already, chances are this one is custom as well, so knowing the exact part number wouldn’t add a lot of extra info. We can see one differential pair going from the analog front-end into this IC and a second one going from this IC to the next one, an ADG286D. National Semi ADG286D Mystery Chip Another custom chip with unknown functionality. Motorola MC10319DW 8-bit 25 MHz A/D Converter Finally, an off-the-shelf device! But why is it only rated for 25MHz? National Semi ADG303 - A Custom Memory Controller Chip It receives the four 8-bit lanes from the four ADCs on one side and connects to four SRAMs on the other. 4 Alliance AS7C256-15JC SRAMs Each memory has a capacity of 32KB and a 15ns access time, which allows for a maximum clock of 66 MHz. The TDS 684B supports waveform traces of 15k points, so they either only use half of the available capacity or they use some kind of double-buffering scheme. There are four unpopulated memory footprints. In one of my TDS 420A blog posts, I extend the waveform memory by soldering in extra SRAM chips. I’m not aware of a TDS 684B option for additional memory, so I’m not optimistic about the ability to expand its memory. There’s also no such grayed-out option in the acquisition menu. When googling for “ADG286D”, I got my answer when I stumbled on this comment on BlueSky which speculates that it’s an analog memory, probably some kind of CCD FIFO. Analog values are captured at a rate of up to 5 GHz and then shifted out at a much lower speed and fed into the ADC. I later found a few other comments that confirm this theory. Measuring Along the Signal Path Let’s verify this by measuring a few signals on the board with a different scope. The ADC input pins are large enough to attach a Tektronix logic analyzer probe: ADC sampling the signal With a 1 MHz signal and using a 100Msps sample rate, the input to the ADC looks like this: The input to the ADC is clearly chopped into discrete samples, with a new sample every 120 ns. We can discern a sine wave in the samples, but there’s a lot of noise on the signal too. Meanwhile the TDS684B CRT shows a nice and clean 1 MHz signal. I haven’t been able to figure out how that’s possible. For some reason, simply touching the clock pin of the ADC with a 1 MOhm oscilloscope probe adds a massive amount of noise to the input signal, but it shows the clock nicely: The ADC clock matches the input signal. It’s indeed 8.33 MHz. Acquistion refresh rate The scope only records in bursts. When recording 500, 1000 or 2500 sample points at 100Msps, it records a new burst every 14ms or 70Hz. When recording 5000 points, the refresh rate drops to 53Hz. For 15000 points, it drops even lower, to 30Hz: Sampling burst duration The duration of a sampling burst is always 2 ms, irrespective of the sample rate of the oscilloscope or the number of points acquired! The combination of a 2 ms burst and 8 MHz sample clock results in 16k samples. So the scope always acquires what’s probably the full contents of the CCD FIFO and throws a large part away when a lower sample length is selected. Here’s the 1 MHz signal sampled at 100 Msps: And here’s the same signal sampled at 5 Gsps: It looks like the signal doesn’t scan out of the CCD memory in the order it was received, hence the signal discontinuity in the middle. Sampling a 1 GHz signal I increased the input signal from 1 MHz to 1 GHz. Here’s the ADC input at 5 Gsps: With a little bit of effort, you can once again imagine a sine wave in those samples. There’s periodicity of 5 samples, as one would expect for a 1 GHz to 5 Gsps ratio. The sample rate is still 8.3 MHz. Sampling a 200 MHz signal I also applied a 200 MHz input signal. The period is now ~22 samples, as expected. 200 MHz is low enough to measure with my 350 MHz bandwidth Siglent oscilloscope. To confirm that the ADG286D chip contains the CCD memory, I measured the signal on one of the differential pins going into that chip: And here it is, a nice 200 MHz signal: A Closer Look at the Noise Issue After initially publishing this blog post, I had a discussion on Discord about the noise issue which made me do a couple more measurements. Input connected to ground Here’s what the ADC input looks like when the input of the scope is connected to ground: 2 major observations: there’s a certain amount of repetitiveness to it. there are these major voltage spikes in between each repetition. They are very faint on the scope shot. Let’s zoom in on that: The spikes are still hard to see so I added the arrows, but look how the sample pattern repeats after each spike! The time delay between each spike is ~23.6 us. With a sample rate of 120ns, that converts into a repetitive pattern of ~195 samples. I don’t know why a pattern of 195 samples exists, but it’s clear that each of those 195 locations have a fixed voltage offset. If the scope measures those offsets during calibration, it can subtract them after measurement and get a clean signal out. 50 kHz square wave Next I applied a 50kHz square wave to the input. This frequency was chosen so that, for the selected sample rate, a single period would cover the 15000 sampling points. 2 more observations: the micro-repetitiveness is still there, irrespective of the voltage offset due to the input signal. That means that subtracting the noise should be fine for different voltage inputs. We don’t see a clean square wave outline. It looks like there’s some kind of address interleaving going on. 50kHz sawtooth wave We can see the interleaving even better when applying a sawtooth wavefrom that covers one burst: Instead of a clean break from high-to-low somewhere in the middle, there is a transition period where you get both high and low values. This confirms that some kind of interleaving is happening. Conclusion The TDS684B captures input signals at high speed in an analog memory and digitizes them at 8 MHz. The single-ended input to the ADC is noisy yet the signal looks clean when displayed on the CRT of the scope, likely because the noise pattern is repetitive and predictable. In addition to noise, there’s also an interleaving pattern during the reading out of the analog FIFO contents. The number of samples digitized is always the same, irrespective of the settings in the horizontal acquisition menu. (Not written by ChatGPT, I just like to use bullet points…)

Introduction The Rohde & Schwarz AMIQ Modulation Generator WinIQSim Software Inside the AMIQ The Signal Generation PCB Analog Signal Generation Architecture: Fixed vs Variable DAC Clock Internal Reference Clock Generation DAC Clock Synthesizer I/Q Output Skew Tuning Variable Gain Amplifier Internal Diagnostics Efficient Distribution of Configuration Signals Conclusion References Introduction Every few months, a local company auctions off all kinds of lab, production and test equipment. I shouldn’t be subscribed to their email list but I am, and that’s one way I end up with more stuff that I don’t really need. During a recent auction, I got my hands on a Rohde & Schwarz AMIQ, an I/Q modulation generator, for a grand total of $45. Add to that another 30% for the auction fee and taxes and you’re still paying much less than what others would pay for a round of golf? But instead of one morning of fun, this thing has the potential to keep me busy for many weekends, so what a deal! (Click to enlarge) A few days after “winning” the auction, I drove to a dark dungeon of a warehouse in San Jose to pick up the loot. The AMIQ has a power on/off button and 3 LEDs and that’s in terms of user interface. There are no dials, there’s no display. So without any other options, I simply powered it up and I was immediately greeted by the ominous clicking of a hard drive. I was right: this thing would keep me entertained for at least a little bit! (Click to enlarge) It took a significant amount of effort to restore the machine back to its working state. I’ll write about that in future blog posts, but let’s start with an overview of the functionality and a teardown of the R&S AMIQ and then deep dive into some of its analog circuits. AMIQ prices on eBay vary wildly, from $129 to $2600 at the time of writing this. Even if you get one of the higher priced ones, you should expect to get a unit that’s close to failing due to leaking capacitors and a flaky harddrive! The Rohde & Schwarz AMIQ Modulation Generator Reduced to an elevator sales pitch, the AMIQ is a 2-channel arbitrary waveform generator (AWG) with a deep sample buffer. That’s it! It has a streaming buffer that feeds samples to 2 14-bit DACs at a sample rate of up to 105MHz. Two output channels I and Q will typically contain quadrature modulation signals that are sent to an RF vector signal generator such as a Rohde & Schwarz SMIQ for the actual high-frequency modulation. In a typical setup, the AMIQ is used to generate the baseband modulated signal and the SMIQ shifts the baseband signal to an RF frequency. Since the AMIQ has no user interface, the waveform data must provided by an external device. This could be a PC that runs R&S WinIQSim software or even the SMIQ itself because it has the ability to control an AMIQ. You can also create your own waveforms and upload them via floppy disk, GPIB or an RS-232 interface using SCPI control commands. Figure 4-1 of the AMIQ Operating Manual has a simplified block diagram. It is pretty straightforward and somewhat similar to the one of my HP 33120A function generator: (Click to enlarge) On the left are 2 blocks that are shared: a clock synthesizer waveform memory And then for each channel: 14-bit D/A converter analog filters output section with amplifier/attenuator differential analog output driver (AMIQ-B2 option) The major blocks are surrounded by a large amount of DACs that are used to control everything from the tuning input of local 10MHz clock oscillator, gain and offset of output signals, clock skew between I and Q signal and much more. You could do some of this with a modern SDR setup, but the specifications of the AMIQ units are dialed up a notch. Both channels are completely symmetrical to avoid modulation errors at the source. If there’s a need to compensate for small delay differences in, for example, the external cables, you can compensate for that by changing the skew between clocks of the output DACs with a precision of 10ps. Similarly, the DAC sample frequency can be programmed with 32-bit precision. (Click to enlarge) In addition to the main I/Q outputs at the front, there are a bunch of secondary input and output signals: 10MHz reference input and output sample clock trigger input marker output external filter loopback output and input bit error rate measurement (BER) connector (AMIQ-B1 option) parallel interface with the digital value of the samples that are sent to the DAC (front panel, AMIQ-B3 option) (Click to enlarge) Above those speciality inputs and outputs are the obligatory GPIB interface and a bunch of generic connectors that look suspiciously like the ones you’d find on an early century PC: PS/2 keyboard and mouse Parallel port RS232 USB There are 3 different AMIQ versions: 1110.2003.02: 4M samples 1110.2003.03: 4M samples 1110.2003.04: 16M samples Mine is an AMIQ-04. WinIQSim Software While it is possible to control the AMIQ over RS-232 or GPIB with your own software, you’d have a hard time matching the features of WinIQSim, a R&S Windows application that supports the AMIQ. (Click to enlarge) With WinIQSim, you can select one of the popular communication protocols from the late nineties and early 2000s, fill in digital framing data, apply all kinds of distortions and interferences, compute the I and Q waveforms and send it to the AMIQ. Some of the supported formats include CDMA2000, 802.11a WLAN, TD-SCDMA and more. But you don’t have to use these official protocols, WinIQSim supports any kind of FSK, QPSK, QAM or other common modulation method. You need a license for some of the communication protocols. The license is linked to the AMIQ device, not the PC, but the license check is pretty naive, and while I haven’t tried it… yet, the EEVblog forum has discussions about how to enable features yourself. My device only came with a license for IS-95 CDMA. Inside the AMIQ It’s trivial to open up an AMIQ: after removing the 4 feet in the back with a regular Philips screwdriver, you can simply slide off the outer case. It has 2 major subsystems: the top contains all the components of a standard PC (Click to enlarge) the bottom has a signal generation PCB (Click to enlarge) The PCB looks incredibly clean and well laid out and I love how they printed the names of different sections on the metal gray shielding plates. We’ll leave the PC system for a future blog post, and focus on the signal generation PCB. The Signal Generation PCB Let’s remove the shielding plates to see what’s underneath. I had to drill out one of the screws that attach the plates because the head was stripped. (Did somebody before me already try to repair it?) (Click to enlarge) The bottom half left and right sections are perfectly symmetrical, as one would expect for a device that has the ability to tune skew mismatches with a 10 ps precision. Annotated, it looks like this: (Click to enlarge) Rohde & Schwarz recently made the terrible decision to lock all their software and manuals behind an approval-only corporate log-in wall, but luckily some of the most important AMIQ assets can be found online elsewhere, including the operating manual and a service manual contains the full schematics! Let’s dig a bit deeper into the various aspects of the design. In what follows I’ll be focusing primarily on analog aspects of the design. This is a very personal choice: not that the digital sections aren’t important, it’s just that, as digital design engineer, they’re not particular interesting to me. By studying the analog sections, I hope to stumble into circuits that I didn’t really know a lot about before. Fantastic Schematics Before digging in for real, a word about the schematics: they are fantastic. Each sub-system has a block diagram that is already pretty detailed, with signal names that match the schematics and test points, often with annotations to indicate the voltage or frequency range. Here’s the block diagram of the reference and DAC clock generation section, for example. Schematic page 5 (Click to enlarge) Signals that come from or go to other pages are fully referenced. Look at the SYN_OUT_CLK signal below: Schematic page 10 (Click to enlarge) The signal is also used on page 6, coordinate 1D and 7B and page 22, coordinate 8A. How cool is that? Signal path test points One of the awesome features of the PCB is the generous amount of test points. We’re not just talking PCB test point against which you can hold your oscilloscope probe or even header pins, though there are plenty of those too, but full on SMB connectors. In addition to these SMD connectors, there are also plenty of jumpers that can be used to interrupt the default signal flow and insert your own test signal instead. Analog Signal Generation Architecture: Fixed vs Variable DAC Clock In the HP 33120A the DAC has a fixed 40 MHz clock. There’s a 16 kB waveform RAM that contains, say, one quarter period of a 100 Hz sine. If you want to send out a sine wave of 200 Hz, instead of sequentially stepping through all the addresses of the waveform RAM, you just skip every other address. One of the benefits of this kind of generation scheme is that you can make do with a fixed frequency analog anti-aliasing filter: the Nyquist frequency is always the same after all. A major disadvantage, however, is that even if the output signal has a bandwidth of only 1MHz, you still need to feed the DAC at the fixed clock rate. You could insert a digital upsampling filter between the waveform memory and the DAC, but that requires significant mathematical DSP fire power, or you’d have to increase the depth of the waveform memory. For an arbitrary waveform generator, it makes more sense to run the DAC at whichever clock speed is sufficient to meet the Nyquist requirement of the desired signal and provide a number of different filtering options. The AMIQ has 4 such options: no filter, a 25 MHz, a 2.5 MHz or a loopback through an external filter. The DAC sample clock range is huge, from 10 Hz all the way to 105 MHz, though specifications are only guaranteed up to 100 MHz. According to the data sheet, the clock frequency can be set with a precision of 10^-7. Internal Reference Clock Generation Like all professional test and measurement equipment, the AMIQ uses a 10MHz reference clock that can come from outside or that can be generated locally with a 10MHz crystal. It’s common for high-end equipment to have an oven controlled crystal oscillator (OCXO), but AMIQ has a lower spec’ed temperature controlled one (TCXO), a Milliren Technologies 453-0210. If we look at the larger reference clock generation block diagram, we can see something slightly unusual: instead of selecting between the internal TCXO output or the external reference clock input, the internal reference clock always comes from the TCXO (green). Schematic page 5 (Click to enlarge) When the internal clock is selected, the TCXO output frequency can be tuned with an analog signal that comes from the VTXCO TUNE DAC (blue), but when the external reference input is active, the TCXO is phase locked to the external clock. You can see the phase comparator and low pass filter in red. The reason for using a PLL with the internal TCXO as the voltage controlled oscillator is probably to ensure that the generated reference clock has the phase noise of the TCXO while tracking the frequency of the external reference clock: a PLL acts as a low-pass filter to the reference clock and as a high-pass filter to the VCO. If the TCXO has a better high frequency phase noise than the external reference clock, that makes sense. This is really out of my wheelhouse, so take all of this with a grain of salt… SYN_REF is the output of the internal reference clock generation unit. DAC Clock Synthesizer The clock synthesizer creates a highly programmable DAC clock from the internal reference clock SYN_REF from previous section. It should come as no surprise that this clock is generated by a PLL as well. Schematic page 5 (Click to enlarge) There are two speciality components in the clock generation path: a Mini-Circuits JTOS-200 VCO an Analog Devices AD9850 DDS Synthesizer (Click to enlarge) The VCO has an operating frequency between 100 and 200 MHz. I don’t know enough about VCOs to give meaningful commentary about the specifications, but based on comparisons with similar components on Digikey, such as this FMVC11009, it’s safe to assume that it’s an expensive and high quality component. The AD9850 is located in the feedback path of the PLL where it acts as a feedback divider with a precision of 32 bits. The signal flow inside the AD9850 is interesting: It works as follows: each clock cycle, the phase of a numerical controlled oscillator (NCO) accumulates with the programmable 32-bit increment. the upper bits of the phase accumulator serve as the address of a sine waveform table. a 10-bit DAC converts the digital sine wave to analog. the analog signal is sent through an external low-pass filter. the output of the low-pass filter goes back into the AD9850 and through a comparator to generate a digital output clock. This thing has its own programmable signal generator! But why the roundtrip from digital to analog back to digital? In theory, the MSB of the NCO could be used as output clock of the clock generator. The problem is that in such a configuration, the length of each clock period toggles between N and N+1 clock cycles, with a ratio so that the average clock period ends up with the desired value. But this creates major spurs in the frequency spectrum of the generated clock. When fed into the phase comparator of a PLL, these frequency spurs can show up as jitter at the output of the PLL and thus into the spectrum of the generated I and Q signals. By converting the signal to analog and using a low pass filter, the spurs can be filtered away. The combination of a low pass filter and a comparator acts as an interpolator: the edges of the generated clock fall somewhere in between N and N+1. Schematic page 21 (Click to enlarge) The steepness of the low pass filter depends on the ratio between the input and the output clock: the lower the ratio, the steeper the filter. There are a bunch of binary clock dividers that make it hard to know the exact ratio, but if the output of the VCO is 100 MHz and the input 10 MHz or less, there is a 10:1 ratio. The AMIQ has a 7th order elliptical low pass filter. I ran a quick simulation in LTspice to check the behavior. (dds_filter.asc source file.) The filter has a cut-off frequency of around 13 MHz: In modern fractional PLLs, instead of using a regular DAC, one could use a high-order sigma-delta unit to create a pulse density modulated output with the noise pushed to higher frequencies and a low pass filter that can be less aggressive. There’s a plenty of literature online about DDS clock generators, some of which I’ve listed in the references at the bottom, but the datasheet of the AD9850 itself is a good start. I/Q Output Skew Tuning Earlier I mentioned the ability to tune the skew between the I and the Q output to compensate for a difference in cable length when connecting the AMIQ to an RF signal generator. While the digital waveform fetching circuit works on one clock, the two clocks that go to the DAC are the ones that can be changed: Here’s what the skewing circuit looks like: Schematic page 10 (Click to enlarge) Starting with input signal DAC_CLK, the goal is to create I_DAC_CLK and Q_DAC_CLK that can be moved up to 1ns ahead or behind the other. Signals can be delayed by sending them through an R/C combo. Since we need a variable delay, we need a way to change the value of either the R or the C. A varactor or varicap diode is exactly that kind of device: its capacitance changes depending on the reverse bias voltage across the diode. Static input signal SKEW_TUNE comes from a DAC. It goes to the cathode of one varicap and the anode of the other. When its voltage increase, the capacitance of those 2 diodes moves in opposite ways, and so does the R/C delay along the I and Q clock path. A BB147 varicap has a capacitance that varies between 2pF and 112pF depending on the bias voltage. Capacitances C168 and C619 prevent the relatively high bias voltages from reaching the digital signal path. Does the circuit work? Absolutely! The scope photos below show the impact of the skewing circuit when dialed to the maximum in both directions: With a 2ns/div horizontal scale, you can see a skew of ~1ns either way. Variable Gain Amplifier The analog signal path starts at the DAC and then goes through the anti-aliasing filters, an output section that does amplification and attenuation, and finally the output connector board. It is common for signal generators to have a fixed gain amplification section and then some selectable fixed attenuation stages: amplifiers with a variable gain and low distortion are hard to design. If you need a signal amplitude that can’t be achieved with one of the fixed attenuation settings, one solution is to multiply the signal before it enters the DAC to reduce the output amplitude, though that’s at the expense of losing some of the dynamic range of the DAC. This is not the case for the AMIQ: while it can use a signal path that only uses fixed amplification and attenuation stages, it also offers the intriguing option to send the analog signal through an analog multiplier stage: If we zoom down from the system diagram to the block diagram, we can see how the multiplier/variable attenuator sits between the filter and the output amplifier, with 2 control input signals: AMPL_CNTRL and OFFSET. This circuit exists twice, of course, for the I and the Q channel. Schematic page 3 (Click to enlarge) Let’s check out the details! The heavy lifting of the variable gain amplifier is performed by an Analog Devices AD835 250 MHz, Voltage Output, 4-Quadrant Multiplier, a part from their Analog Multipliers & Dividers catalog. Schematic page 14 (Click to enlarge) It’s a tiny 8-pin device that calculates W = (X1-X2) * (Y1-Y2) + Z. In low volume, it will set you back $32 on Digikey. In addition to the multiplication, you can set a fixed output gain by feeding back output W through a resistive divider back into Z. For inputs less than 10 MHz, it has a typical harmonic distortion of -70dB. Analog Devices datasheets usually have a Theory of Operation section that explain, well, the underlying theory of operation. The AD835 has such a section as well, but it doesn’t go any further than stating that the multiplier is based on a classic form, having a translinear core, supported by three (X, Y, and Z) linearized voltage-to-current converters, and the load driving output amplifier. I have no clue how the thing works! In the case of the AMIQ, X2 and Y2 are strapped to ground, and there’s no feedback from W back into Z either which reduces the functionality to: W = FILTER_OUT * AMPL_VAR + OFFSET. AMPL_VAR and OFFSET are static analog values that are each created by a 12-bit DAC8143, just like many other analog configuration signals. It’s almost shame that this powerful little device is asked to perform such a basic operation. While researching the AD835, somebody pointed out the AD8310, another interesting speciality chip from Analog Devices. It’s a DC to 440 MHz, 95dB logarithmic amplifier, a converter from linear to logarithmic scale basically. Discovering little gems like this is why I love studying schematics of complex devices. Internal Diagnostics The AMIQ has a set of 15 internal analog signal to monitor the health of the device. Various meaningful signals are gathered all around the signal generation board and measured at power up to give you that dreaded Diagnostic Check Fail message. Schematic page 27 (Click to enlarge) The circuit itself is straightforward: 2 8-to-1 analog multiplexers feed into a CS5507AS 16-bit AD converter. It can only do 100 samples per second, but that’s sufficient to measure a bunch of mostly static signals. Like many other devices, the measured value is sent serially to one of the FPGAs. The ADC needs a 2.5V reference voltage. It’s funny that this reference voltage also goes to the analog multiplexers as one one of the diagnostic signals. One wonders what the ADC returns as a result when it tries to convert the output of a broken reference voltage generator. There are a bunch of circuits in the AMIQ whose only purpose is generating diagnostic signals. Here’s a good example of that: Schematic page 14 (Click to enlarge) I_OUT_DIAG and Q_OUT_DIAG are the outputs after the attenuator that will eventually end up at the connectors. The circuit in the top red square is a signal peak detector, similar to what you’d find in the rectifier of a power supply. It allows the AMIQ to track the output level of signals with a frequency that is way higher than the sample rate of the ADC. The circuit in the red rectangle below performs a digital XOR on the analog I and the Q signals and then sends it through a simple R/C low pass filter. I think that it allows the AMIQ to check that the phase difference between the I and Q channel is sensible when applying a test signal during power-on self-test. Efficient Distribution of Configuration Signals The AMIQ signal board has hundreds of configuration bits: there are the obvious enable/disable or selection bits, such as those to select between different output filters, but the lion’s share are used to set the output value of many 12-bit DACs. Instead of using parallel busses that fan out from an FPGA, the AMIQ has a serial configuration scan chain. Discrete 8-bit 74HCT4094 shift-and-store registers are located all over the design for the digital configuration bits. The DAC8134 devices have their own built-in shift register and are part of the same scan chain. Schematic page 19 (Click to enlarge) The schematic above is an example of that. The red scan chain data input goes through the VTCXO_TUNE DAC, then the 74HCT4094 after which it exist to some other page of the schematics. Conclusion Around the late nineties, test equipment companies started to stop adding schematics to their service manuals, but the R&S AMIQ is a nice exception to that. And while the device already has a bunch of FPGAs, most of the components are off-the-shelf single-function components. Thanks to that, the AMIQ is an excellent candidate for a deep dive: all the information is there, you need to just spend a bit of effort to go through things. I had a ton of fun figuring out how things worked. References Rohde & Schwarz documents R&S - IQ Modulation Generator AMIQ R&S - AMIQ Datasheet R&S - AMIQ Operating Manual R&S - AMIQ Service Manual with schematic Various application notes R&S - Floppy Disk Control of the I/Q Modulation Generator AMIQ R&S - Software WinIQSIM for Calculating I/Q Signals for Modulation Generator R&S AMIQ R&S - Creating Test Signals for Bluetooth with AMIQ / WinIQSIM and SMIQ R&S - WCDMA Signal Generator Solutions R&S - Golden devices: ideal path or detour? R&S - Demonstration of BER Test with AMIQ controlled by WinIQSIM Other AMIQ content on the web zw-ix has a blog - Getting an Rohde Schwarz AMIQ up and running zw-ix has a blog - Connecting a Rohde Schwarz AMIQ to a SMIQ04 Bosco tweets DDS Clock Synthesis MT-085 Tutorial: Fundamentals of Direct Digital Synthesis (DDS) How to Predict the Frequency and Magnitude of the Primary Phase Truncation Spur in the Output Spectrum of a Direct Digital Synthesizer (DDS) Related content The Signal Path - Teardown, Repair & Analysis of a Rohde & Schwarz AFQ100A I/Q (ARB) Modulation Generator